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Introduction
The power inductor is the energy storage element at the heart of every switching regulator, buck converter, boost converter, and multiphase VRM. Its electrical parameters determine converter efficiency, output voltage ripple, thermal performance, and EMI signature. Its physical package determines how tightly it can be integrated into a high-density PCB layout. And yet, power inductors are frequently selected by a single criterion—inductance value—while the parameters that actually determine real-world performance—saturation current, DCR, core material, and package type—are treated as secondary details.
This guide covers every parameter that matters when selecting a power inductor for a switching power supply, VRM, or any other application that requires high-current energy storage on a PCB. It is written for hardware engineers who need to make the right selection on the first pass, without discovering at board bring-up that the inductor saturates under load, runs excessively hot, or radiates EMI into adjacent circuits.
In a synchronous buck converter, the inductor alternately stores energy from the input during the switch-on phase and releases that energy to the output during the switch-off phase. The inductor current ramps up linearly during the on-phase (governed by VL = L × di/dt) and ramps down during the off-phase. The peak-to-peak amplitude of this current ripple (ΔIL) is one of the central design trade-offs:
The output ripple voltage ΔVout ≈ ΔIL × ESRcout + ΔIL / (8 × fsw × Cout), where fsw is switching frequency and Cout is output capacitance. Understanding this relationship is essential before selecting inductance value, because the interaction between the inductor and the output capacitor determines whether the converter's output meets its ripple specification. For the decoupling capacitor role in this system, the decoupling capacitor placement guide provides the complementary context.
A power inductor datasheet typically specifies the following parameters. All of them matter; none can be ignored in a well-designed converter.
| Parameter | Symbol | Definition | Design Impact |
|---|---|---|---|
| Inductance | L | Inductance value in μH or nH, measured at specified frequency and current | Determines ripple current ΔIL and transient response speed |
| Saturation current | Isat | DC current at which inductance drops by a specified percentage (typically 20% or 30%) | Sets the maximum peak current the inductor can handle without excessive inductance drop |
| RMS current rating | Irms | Current at which the inductor reaches a specified temperature rise (typically 40°C) | Sets the continuous current capability limited by thermal dissipation in DCR |
| DC resistance | DCR | Resistance of the copper winding at DC, in mΩ | Determines conduction losses (PDCR = I2 × DCR) and self-heating |
| Self-resonant frequency | SRF | Frequency at which the parasitic capacitance resonates with the inductance | Above SRF the component behaves as a capacitor; switching frequency must be well below SRF |
| Q factor | Q | Ratio of reactive impedance to resistive impedance at a given frequency | Higher Q = lower loss; critical for RF inductors; less emphasized for power inductors |
The inductance value is typically determined by the target ripple current as a fraction of maximum load current. The standard design guideline is to select inductance such that ΔIL = 20–40% of Iout(max). This range balances ripple performance against inductor size and transient response speed.
The required inductance for a synchronous buck converter is:
L = (Vin − Vout) × Vout / (Vin × fsw × ΔIL)
Where:
Example calculation: A 5 V to 1.8 V buck converter at 1 MHz switching frequency with Iout(max) = 10 A and target ΔIL = 30% × 10 A = 3 A:
L = (5 − 1.8) × 1.8 / (5 × 1,000,000 × 3) = 5.76 / 15,000,000 ≈ 384 nH
In practice, the nearest standard value above 384 nH would be selected—typically 390 nH or 470 nH. Always round up rather than down to keep ripple below the target.
Higher switching frequencies allow smaller inductance values for the same ripple current, which enables physically smaller inductors. This is why modern DC-DC converters operating at 1–5 MHz use inductors in the 100–470 nH range, while older designs at 200–300 kHz required 4.7–22 μH for the same application.
Saturation current is the most critical parameter for power inductor selection and the most commonly misunderstood. When a ferromagnetic core carries current above its saturation threshold, the magnetic flux density reaches the saturation value of the core material and the effective permeability drops dramatically. This causes the inductance to decrease sharply—in some cases by 50–80% at the saturation current rating.
If the inductor saturates during converter operation, several problems follow simultaneously:
The Isat specification is not uniform across manufacturers. Some specify Isat as the current at 20% inductance drop; others use 30% or 35%. This makes direct comparison between manufacturers using their headline Isat number misleading. Always check the specification definition in the datasheet and compare at the same percentage drop.
Selection rule: The inductor's Isat must exceed the maximum peak inductor current, which is Iout(max) + ΔIL/2. With an appropriate derating factor (typically 20–30% for the inductance drop at Isat), the selection criterion becomes:
Isat ≥ (Iout(max) + ΔIL/2) / 0.75
For the 10 A converter example above with 3 A ripple: Isat ≥ (10 + 1.5) / 0.75 ≈ 15.3 A. An inductor rated for Isat = 15 A would be at its limit; 18–20 A is a safer selection.
The saturation behavior of composite alloy core inductors differs from ferrite: composite cores exhibit a softer, more gradual saturation characteristic (inductance decreases smoothly with current) while ferrite cores saturate more abruptly. Soft saturation provides some warning before complete saturation; hard saturation can produce near-instantaneous collapse of inductance.
While Isat sets the peak current limit, Irms sets the continuous thermal limit. Power is dissipated in the winding resistance as P = Irms2 × DCR. When the continuous RMS current through the inductor heats the winding above a specified temperature rise (typically 20°C or 40°C above ambient), the inductor is operating at or above its Irms rating.
The RMS current in a buck converter is approximately:
IL(rms) = √(Iout2 + (ΔIL)2 / 12)
For most practical designs where ΔIL < 30% of Iout, IL(rms) is very close to Iout. The Irms rating must exceed IL(rms) with a thermal derating margin that accounts for the PCB ambient temperature and the self-heating of nearby components. A conservative practice is to select Irms ≥ 1.25 × IL(rms).
For multiphase VRM designs (as used in CPU and GPU power delivery), the per-phase inductor carries Iload / n, where n is the number of phases, allowing a much smaller per-inductor current rating. This is one of the primary advantages of multiphase designs for high-current loads.
DCR is the copper winding resistance at DC. It is the dominant source of conduction loss in a power inductor at low-to-moderate switching frequencies. The power dissipated in the DCR is Ploss = Irms2 × DCR, and this heat must be removed from the inductor to maintain junction temperature within rating.
The trade-off between DCR and Isat / Irms is fundamental to inductor design: lower DCR requires thicker wire, which limits the number of turns that fit in a given core volume, reducing inductance and Isat for the same package size. Higher DCR allows more turns in the same volume (thinner wire), increasing inductance and Isat but at the cost of higher conduction losses.
| DCR Level | Typical Range | Suitable Applications | Trade-offs |
|---|---|---|---|
| Ultra-low DCR | < 2 mΩ | CPU/GPU VRM, AI accelerator PDN, high-current server power | Large package; limits maximum inductance for given package |
| Low DCR | 2–10 mΩ | High-efficiency industrial power, telecom, general server | Good balance of efficiency and size |
| Moderate DCR | 10–50 mΩ | Consumer electronics, portable devices, IoT | Smaller package; acceptable efficiency at lower currents |
| High DCR | > 50 mΩ | Low-current auxiliary supplies, signal filtering | Highest inductance per package; poor efficiency at high current |
At switching frequencies above approximately 1 MHz, AC resistance (due to skin effect and proximity effect in the winding) becomes significant and may add 30–100% to the effective resistance compared to DCR alone. High-frequency core losses (hysteresis and eddy current losses) also increase with switching frequency. For converters above 2–3 MHz, the manufacturer's published AC loss data or core loss curves should be used in addition to the DCR specification.
Every inductor has a parasitic distributed capacitance from winding turn-to-turn and between the winding and the core/shield. At the SRF, this capacitance resonates with the inductance, producing a parallel resonance where the component presents very high impedance. Above the SRF, the component behaves as a capacitor rather than an inductor.
For power inductor applications, the switching frequency must be well below the SRF—typically by a factor of at least 10:1 for reliable inductor behavior. A power inductor with SRF of 100 MHz is suitable for converters switching up to approximately 10 MHz; at higher frequencies, the SRF must increase proportionally.
SRF is inversely related to inductance: larger inductance values have more winding turns and more distributed capacitance, resulting in lower SRF. Very high inductance values (10–100 μH) may have SRF in the 10–50 MHz range, limiting their use to lower-frequency converters. Smaller inductance values (100–470 nH) typically have SRF above 200–500 MHz, suitable for high-frequency switching at 1–20 MHz.
The core material determines the inductor's saturation characteristics, temperature stability, frequency performance, and cost.
Ferrite (MnZn or NiZn): The most common core material for power inductors in switching converters. High permeability enables high inductance in small packages. Relatively abrupt saturation characteristic—inductance drops sharply when saturation is reached. Good high-frequency performance (MnZn suitable to ~5 MHz; NiZn suitable to hundreds of MHz). Poor mechanical stress resistance (brittle ceramic). Most shielded SMD power inductors use ferrite cores.
Composite alloy powder (metal alloy powder): Increasingly common in high-current, high-frequency applications. Soft saturation characteristic—inductance decreases gradually as current increases, rather than the abrupt cliff of ferrite saturation. Better performance under mechanical vibration and thermal shock. Higher saturation flux density than ferrite, enabling higher Isat in smaller packages. Lower permeability than ferrite requires more turns for the same inductance, increasing DCR slightly. Used in premium VRM inductors for CPUs and GPUs where the soft saturation behavior prevents catastrophic inductance collapse during current transients.
Iron powder (MPP, Kool Mμ, High Flux): Gapped ferrite equivalents with distributed air gaps in the core material. Very soft saturation characteristic; inductance decreases very gradually with current. High saturation flux density. Primarily used in toroidal inductor forms for output filter chokes in power supplies, EMI filters, and PFC inductors. Less common in SMD switching converter inductors.
| Core Type | Saturation Characteristic | Frequency Range | Relative DCR | Primary Applications |
|---|---|---|---|---|
| MnZn Ferrite | Abrupt (hard) | 100 kHz – 5 MHz | Low | General-purpose DC-DC, power supply output filters |
| NiZn Ferrite | Abrupt | 1 MHz – 200 MHz | Low | High-frequency DC-DC, RF chokes |
| Composite alloy (metal powder) | Soft (gradual) | 100 kHz – 20 MHz | Moderate | CPU/GPU VRM, AI server PDN, automotive |
| Iron powder (MPP, Kool Mμ) | Very soft | 10 kHz – 1 MHz | Low to moderate | PFC chokes, output filter chokes, EMI filters |
| Amorphous/Nanocrystalline | Soft | 1 kHz – 500 kHz | Low | High-power PFC, transformers, common mode chokes |
Shielded inductors: The winding is completely enclosed in a ferrite or metal shield, preventing external magnetic flux leakage. Nearly eliminates radiated EMI to adjacent components. The closed magnetic path also improves inductance stability. Higher cost and slightly larger height than equivalent unshielded types. Essential in high-density PCBs where sensitive analog or RF circuits are present within a few millimeters. Preferred for professional and industrial designs.
Semi-shielded inductors: Partial shielding—the core provides containment in some directions but not all. A compromise between the EMI performance of shielded types and the cost/size of unshielded types. Suitable for medium-density designs without extremely sensitive adjacent circuits.
Unshielded inductors (open construction): Drum core or bobbin construction with no external shield. Lowest cost and smallest height for a given inductance/current rating. Significant magnetic flux leakage that can couple into adjacent traces, ICs, or other inductors. Requires larger keep-out zones and careful PCB orientation to minimize EMI coupling. Generally limited to less density-sensitive designs, power supplies with ample spacing from sensitive circuits, or cost-optimized consumer electronics.
The PCB layout implications of each type differ significantly. Unshielded inductors should be oriented so their winding axis is perpendicular to adjacent sensitive signal traces, and the same orientation should be maintained for all unshielded inductors in the same power supply to prevent flux coupling between phases. Shielded inductors can be placed with fewer orientation constraints, but still benefit from a copper ground plane beneath them to reduce fringe fields.
Power inductor packages are designated by dimensions (length × width × height in mm) rather than the EIA package codes used for resistors and capacitors. Common package designations and their approximate current capability:
| Package (L × W mm) | Typical Height Range | Typical Isat Range | Typical Inductance Range | Common Applications |
|---|---|---|---|---|
| 2.0 × 2.0 | 0.8–1.2 mm | 1–5 A | 47 nH – 4.7 μH | IoT, wearables, low-current MCU supply |
| 3.0 × 3.0 | 1.0–1.5 mm | 2–8 A | 100 nH – 10 μH | Mobile SoC, display driver supply |
| 4.0 × 4.0 | 1.2–2.0 mm | 5–15 A | 220 nH – 22 μH | Laptop CPU supply, FPGA core, DDR supply |
| 5.0 × 5.0 | 1.5–2.5 mm | 10–25 A | 330 nH – 47 μH | Desktop CPU, server individual phase |
| 6.0 × 6.0 | 2.0–4.0 mm | 15–40 A | 470 nH – 100 μH | Server VRM phase, point-of-load for GPU |
| 7.0 × 7.0 and larger | 3.0–6.0 mm | 30–100 A+ | 1 μH – 220 μH | High-current server PDN, power module |
Height (Z dimension) is a critical mechanical constraint in low-profile designs. Laptop motherboards and thin server blades may have maximum component height limits of 1.5–2.0 mm. These constraints force selection of the flattest inductor package available at the required current rating, often at higher cost or with a compromise on DCR.
The following compares representative inductors from major manufacturers for a 5 V to 1.8 V / 10 A / 1 MHz buck converter requiring approximately 470 nH inductance with Isat ≥ 15 A:
| Manufacturer / Part | Inductance | Isat (−30%) | Irms (ΔT 40°C) | DCR (typ.) | Package | Height | Core Type |
|---|---|---|---|---|---|---|---|
| TDK VLS5045EX | 470 nH | 16.5 A | 11.0 A | 4.2 mΩ | 5.0 × 5.0 mm | 4.5 mm | Ferrite (shielded) |
| Murata LQM44PN | 470 nH | 18 A | 12 A | 3.8 mΩ | 4.4 × 4.4 mm | 3.0 mm | Metal composite (shielded) |
| Bourns SRR5028 | 470 nH | 14 A | 9.8 A | 6.0 mΩ | 5.0 × 5.0 mm | 2.8 mm | Ferrite (shielded) |
| Vishay IHLP5050 | 470 nH | 20 A | 13 A | 3.5 mΩ | 5.0 × 5.0 mm | 5.0 mm | Metal composite (shielded) |
| Taiyo Yuden MCOILPJ | 470 nH | 15 A | 10.5 A | 4.8 mΩ | 4.5 × 4.5 mm | 3.2 mm | Metal composite (shielded) |
Note: Values are illustrative; always verify against current manufacturer datasheets as specifications change with product revisions.
Power inductor placement and routing directly affect converter efficiency, EMI performance, and thermal management. The following rules apply to most DC-DC converter designs:
Rule 1: Keep the switching loop as small as possible. The high-frequency switching loop consists of the input capacitor, the high-side switch, the low-side switch (or diode), and the inductor. This loop carries fast-switching current and is the primary source of conducted and radiated EMI. Place the input capacitors as close as possible to the switch node, and minimize the PCB trace area enclosed by this loop.
Rule 2: Place the inductor close to the switch node output. The trace from the switch output (SW pin) to the inductor should be as short and wide as possible. This trace carries high-frequency switching transients and benefits from minimum inductance (achieved by shortening and widening the trace) and minimum radiation area.
Rule 3: Do not place sensitive analog signals beneath or adjacent to unshielded inductors. Unshielded inductors radiate significant magnetic flux, which induces voltages in nearby conductors. Clock signals, ADC inputs, PLL supplies, and precision reference voltages should be routed at least 3–5 mm from unshielded inductors. Shielded inductors have much lower fringe fields and allow closer placement of sensitive signals.
Rule 4: Orient inductor winding axes to minimize coupling. If multiple inductors are placed near each other (multiphase VRM), orient their winding axes perpendicular to each other (90° rotation) to minimize mutual inductance between phases. Two parallel-oriented inductor windings in close proximity can couple enough energy to significantly increase ripple on both phases.
Rule 5: Provide thermal relief and copper spreading under the inductor. High-current inductors dissipate significant heat through their winding resistance. A copper thermal pad on the PCB beneath the inductor, connected to a ground plane through multiple vias, provides a heat path from the inductor body to the copper plane mass. For inductors dissipating more than 0.5 W, thermal simulation should verify that junction temperature stays within rating.
Rule 6: Avoid placing inductors directly above or below high-speed signal layers. Inductor flux penetrates the PCB stack and can couple into inner signal layers. For multilayer boards, place inductors over ground plane layers rather than signal routing layers, and avoid routing sensitive high-speed traces directly beneath high-current inductors.
Rule 7: Use wide traces on the inductor output side. The trace from the inductor output to the output capacitors and load carries the full DC load current. Size this trace for < 10–20 mV of voltage drop at maximum load current, which typically requires 2–5 mm trace width at 1 oz copper for 10 A loads, or copper fills for high-current applications.
| Application | Switching Freq. | Output Current | Recommended Inductance | Core Type | Package | Key Priority |
|---|---|---|---|---|---|---|
| IoT / low-power MCU (3.3 V out) | 1–3 MHz | 0.5–2 A | 2.2–4.7 μH | Ferrite (shielded) | 2.0 × 2.0 or 3.0 × 3.0 | Size, cost |
| Smartphone / tablet SoC supply | 2–4 MHz | 2–6 A | 470 nH – 2.2 μH | Metal composite | 2.5 × 2.5 or 3.0 × 3.0 | Height, efficiency |
| Laptop CPU core supply | 400 kHz – 1 MHz | 5–15 A per phase | 330 nH – 1 μH | Metal composite | 4.0 × 4.0 or 5.0 × 5.0 | Soft saturation, low DCR |
| Server CPU VRM (multiphase) | 300–800 kHz per phase | 10–30 A per phase | 150–560 nH | Metal composite | 6.0 × 6.0 or 7.0 × 7.0 | Ultra-low DCR, high Isat |
| GPU / AI accelerator VCORE VRM | 300–600 kHz per phase | 20–80 A per phase | 100–330 nH | Metal composite (low profile) | 7.0 × 7.0 or custom module | Lowest DCR, highest Isat, soft sat. |
| Telecom / networking 48 V to 1 V | 200–500 kHz | 20–50 A | 1–4.7 μH | Ferrite or composite | 8.0 × 8.0 or larger | Efficiency, thermal |
| Automotive 12 V to 5 V (BCM/ECU) | 400 kHz – 2 MHz | 1–5 A | 1–10 μH | Ferrite (shielded, AEC-Q200) | 4.0 × 4.0 or 5.0 × 5.0 | AEC-Q200 qual., vibration resistance |
What is the difference between Isat and Irms?
Isat is the current at which the core saturates and inductance drops by a defined percentage (typically 20–30%). It sets the peak current limit based on magnetic behavior of the core. Irms is the continuous current at which the inductor reaches a defined temperature rise (typically 40°C) due to power dissipation in the winding resistance (DCR). In most designs, you must verify that both limits are satisfied: Isat is not exceeded by the peak inductor current (Iout + ΔIL/2), and Irms is not exceeded by the RMS inductor current (approximately equal to Iout for typical ripple levels).
Can I use any inductor for a power converter as long as the inductance value matches?
No. Inductance value alone is insufficient. An RF inductor with the correct inductance but an Isat rating of 0.1 A will saturate immediately at any appreciable power supply current, causing converter malfunction or damage. A power inductor must be selected for inductance, Isat, Irms, DCR, SRF, and core type simultaneously. Using an undersized inductor in a power converter is one of the most common causes of prototype power supply failure.
Why do some inductors list two Isat values?
Some manufacturers specify Isat at two inductance drop levels—for example, Isat at −20% inductance and Isat at −30% inductance. The −20% figure is more conservative and represents the onset of significant saturation; the −30% figure is the traditional industry-standard definition. When comparing inductors from different manufacturers, verify which percentage drop applies to each datasheet to make an apples-to-apples comparison. Always design to the more conservative −20% figure when headroom exists, as operation beyond −20% inductance drop increases ripple current significantly.
Should I always use a shielded inductor for power supply design?
Shielded inductors are strongly recommended for any design that includes sensitive analog circuits, RF components, clock circuitry, or precision measurement functions within 5–10 mm of the power supply. The higher cost of shielded inductors is almost always justified when compared to the board space, design time, and PCB respins needed to debug EMI problems caused by unshielded inductors. For simple, cost-optimized designs with no sensitive neighboring circuitry, unshielded inductors are acceptable if placed with adequate spacing and proper orientation.
How does switching frequency affect inductor selection?
Higher switching frequency allows smaller inductance for the same ripple current (because the inductor operates over shorter on-time intervals), which permits physically smaller inductor packages. However, higher frequency increases core losses (hysteresis and eddy current losses scale with frequency), requiring core materials optimized for high-frequency operation (NiZn ferrite or metal composite) rather than standard MnZn ferrite. The inductor's SRF must also be well above the switching frequency. For converters above 2 MHz, verify the core loss data in the datasheet or manufacturer's simulation tools to ensure efficiency targets can be met.
Correct power inductor selection is only effective when the PCB layout delivers the tight switch loop geometry, adequate copper cross-section, and thermal management required for efficient converter operation. NextPCB provides thick copper PCB fabrication for high-current power delivery traces, SMT assembly with automated placement of power inductors and passive components, and full PCBA capabilities from single prototypes to production volumes.
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